Amplifier providing power recovery from a narrow-band antenna

ABSTRACT

A method, amplifier and system are provided for enabling power recovery from a narrow-band antenna when a signal having bandwidth exceeding that of the antenna is utilized. The amplifier provides amplification of a source signal to the antenna and recovery of power stored in the antenna during periods when the impedance of the antenna is negative to enable reverse current through the amplifier to a direct current (DC) power source.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a divisional of U.S. patent application Ser. No.12/470,210 filed on May 21, 2009 and issued as U.S. Pat. No. 8,140,035on Mar. 20, 2012, which further claims benefit of U.S. ProvisionalPatent Application No. 61/054,831 filed on May 21, 2008 and U.S.Provisional Patent Application No. 61/054,836 filed on May 21, 2008. Thedisclosure of each foregoing patent application is hereby incorporatedherein in its entirety, for all purposes.

TECHNICAL FIELD

The invention relates to radio frequency signal transmission amplifiers.More specifically, it relates to amplifiers in transmitters where theantenna has a relatively short electrical length compared to thetransmitted wavelength.

BACKGROUND

In recent years, traditional radio-navigational systems, such as Loran(long range navigation), have been slowly replaced or relegated to abackup role for more accurate satellite navigational systems, such asGPS (global positioning system). However, complete replacement ofradio-navigational systems with GPS has not occurred thus far, due tosome of the shortcomings of GPS.

In fact, there has been a renewed interest in systems such as Loran toprovide backup for GPS systems, in the event of failure of the GPSsystems. Accordingly, Loran systems are being studied to determinewhether they can be updated to provide a reliable backup system for GPS.

Loran antennas used in most transmission sites are characterized byhaving relatively short electrical lengths compared to the transmittedwavelength. The antenna, highly capacitive due to the short electricallength, is normally series resonated with a loading inductor to minimizethe reactance at the center frequency.

The resulting tuned circuit has a very narrow bandwidth with a qualityfactor (Q) typically in the range of 20 to 60.

The ideal transmitted Loran signal has a bandwidth that considerablyexceeds the bandwidth of the transmission antenna. It is normally arequirement for any transmitter system that the bandwidth of the antennaexceeds the bandwidth of the transmitted signal. In the case of Loran,the antenna bandwidth deficit makes the antenna unsuitable for a typicaltransmitter, such as a Long Wave or Medium Wave Amplitude Modulation(AM) simply modified for operation at 100 kHz.

AM broadcasting transmitters are designed to operate into constantimpedance, typically 50 ohms. The concept of impedance implies a steadystate relationship between the voltages and currents in the amplifiersand the antenna. With Loran, no such steady state relationship exists.The instantaneous impedance of the antenna (the ratio of voltage tocurrent at one instant in time) varies throughout the pulse from a verylarge level initially to close to the steady state base impedance nearthe pulse peak, then decreasing and finally becoming negative during thepulse tail. When the impedance is negative, power is flowing out of theantenna back to the transmitter. The negative power flow necessitatesthe use of “tail biter” circuitry, currently in use in Lorantransmitters.

Typical amplifiers are designed for operation when the load is thoughtof as a resistor meaning that the signal bandwidth is less than theantenna bandwidth and the induced current waveform directly follows thevoltage waveform. AM broadcast transmitters use passive LC impedancematching and combining circuitry to match the antenna impedance to theoptimum load impedance for the radio frequency amplifiers. As a resultof the considerable change in the antenna impedance during the Loranpulse, passive LC impedance matching networks are not ideal for thissituation.

Accordingly, there is a need to develop a more efficient amplifier thatcan be used to transmit a signal that is larger than the bandwidth of anarrow-band antenna.

BRIEF DESCRIPTION OF THE DRAWINGS

Further features, aspects, and advantages of the present invention willbecome apparent from the following detailed description, taken incombination with the appended drawings, in which:

FIG. 1 is a representation of a radio frequency transmitter;

FIG. 2 is a graph representing a Loran pulse in the time domain;

FIG. 3A is a graph of a typical narrow band antenna magnitude response;

FIG. 3B is a graph representing a Loran pulse spectrum;

FIG. 4A is a graph of the Loran pulse envelope, the voltage enveloperequired from the transmitter, which is equalized, and the instantaneousimpedance;

FIG. 4B is a graph showing antenna power flow and efficiency limit;

FIG. 5 is a representation of a step modulation system;

FIG. 6 is a representation of an amplifier for use with a narrow-bandantenna;

FIGS. 7A to 7G are representations of the states of the amplifier;

FIG. 8 is a graph representing combined amplifier output voltagewaveform for Loran pulse; and

FIG. 9 is a method of controlling an amplifier for providing powerrecovery.

It will be noted that throughout the appended drawings, like featuresare identified by like reference numerals.

DETAILED DESCRIPTION

The present disclosure provides an amplitude modulated amplifier forLoran signal transmission. Also disclosed is a system for Loran signaltransmission using the amplitude modulated amplifier. Embodiments of thepresent invention are described below, by way of example only, withreference to FIGS. 1-9.

In accordance with an aspect of the disclosure there is provided amethod for using an amplifier to recover stored power from an antennahaving a short electrical length compared to a wavelength of atransmitted signal. The method includes the steps of: (A) determining animpedance state of the antenna receiving a source signal fortransmission; (B) enabling an amplifier to operate in a driving statewhen the impedance state is positive to modulate the source signal goingto the antenna; and (C) enabling the amplifier to operate in a dampingstate when the impedance state of the antenna is negative, wherein thedamping state provides current from the antenna to charge a directcurrent (DC) power source from power stored in the antenna.

In accordance with an aspect of the disclosure there is provided anamplitude modulated amplifier for amplifying a radio frequency (RF)signal. The amplifier includes: (A) four selectably controllabletransistors arranged to receive power from a direct current (DC) powersupply to amplify the signal, the transistors configured in a bridgeconfiguration with a first and a second transistor coupled between theDC power supply and an antenna having a short electrical length comparedto a wavelength of a transmitted signal, and with a third and a fourthtransistor coupled between the antenna and ground in an ‘H’configuration, each transistor having a gate input to determine thestate of the transistor between an on state and an off state; (B) fourdiodes each associated with one of the four transistors and forwardbiased from a source to a drain of each transistor, forming a bridgerectifier; and (C) drive logic for controlling each of the fourtransistors from an open state to a closed amplifying state based onexternal input, wherein the drive logic is selectable from a drivingstate to a damping state of the amplifier by switching transistors torecover power from the antenna when an impedance state of the antenna isnegative, and to deliver the recovered power to the DC power supply.

In accordance with an aspect of the disclosure there is provided asystem for transmitting radio frequency signals into an antenna having ashort electrical length compared to the wavelength of a transmittedsignal. The system includes: (A) a signal source providing a basebandsource signal to be amplified and modulated before transmission; (B) anexciter module for up-converting the source signal based on a frequencyof a received RF source; (C) an amplification module comprising one ormore amplifier, each amplifier comprising: (i) four selectablycontrollable transistors arranged to receive power from a direct current(DC) power supply to amplify the up-converted source signal, thetransistors configured in a bridge configuration with a first and asecond transistor coupled between the DC power supply and an antenna,and a third and fourth transistors coupled between the antenna andground in an ‘H’ configuration, each transistor having a gate input todetermine the state of the transistor between an on state and an offstate; (ii) four diodes each associated with one of the four transistorsand forward biased from a source to a drain of each transistor, forminga bridge rectifier; and (iii) drive logic for controlling each of thefour transistors from an open state to a closed amplifying state basedon external input, wherein the drive logic is selectable from an activestate to a damping state of the amplifier by switching transistors torecover power from the antenna when an impedance state of the antenna isnegative, and to deliver the recovered power to the DC power supply; (D)a power source for delivering DC power to the amplification module orreceiving and storing power from the amplification module; (E) anantenna coupled to the amplification module, the antenna having a highlycapacitive characteristic wherein the signal wavelength is less than theantenna wavelength, wherein current is delivered from the amplificationmodule to the antenna when in the driving state and impedance ispositive, and current is received from the antenna and delivered to theDC power source in the damping state when impedance is negative; and (E)a controller coupled to or integrated with the exciter module, forcontrolling the operation of the one or more amplifiers via the drivelogic, wherein a subset of the one or more amplifiers is switchedbetween the driving state or damping state, wherein any remainingamplifier(s) of the one or more amplifiers are placed in an inactivemode, to control the modulation of the transmitted signal.

In very general terms, the present disclosure provides a modifiedamplitude modulated amplifier and associated system and method that isdesigned to accommodate and essentially capture energy flowing from anarrow-band antenna during the negative power flow phase of a signalsuch as in a Loran pulse.

A suitable transmitter, as shown in FIG. 1 includes an input source 100,an exciter/modulator 102, an RF source 104, a controller 114, one ormore amplifiers 106, a combiner 108, a harmonic filter 110 and a powersource 118. An antenna 112 is connected to the transmitter to broadcastthe signal.

The input source 100 provides an input signal that is an inphase andquadrature (I, Q) description of the baseband complex envelope. Theinput signal operates through a digital input that carries a basebandrepresentation of the ideal signal to be transmitted. This input wouldbe used when the transmitter operates in the continuous mode.Alternately the transmitter may operate in a pulsed mode where theprimary input is used to trigger a pulse from the transmitter and allpulses are the same. The pulse shape is stored in the transmitterdefining the ideal voltage waveform for a single pulse. A DSP basedequalizer may be used to determine the necessary amplifier voltagewaveform from the desired antenna current waveform when operating in thecontinuous mode. In the pulse mode the ideal voltage waveform wouldnormally be determined before being stored in the transmitter.

The Low Frequency (LF) band has a frequency range of 30 kHz to 300 kHz.Loran-C and its more modern derivatives are transmitted in the LF bandat a centre frequency of 100 kHz. In general, the antenna systems usedfor LF transmission are electrically short because wavelengths are verylong and the electrical length of typical antennas is much less than aquarter wave. For Loran, a quarter wave tower would be 750 meters tall.A common antenna used by the United States Coast Guard is the 625 foot(190.5 meters) Top Loaded Monopole (TLM) having an electrical length ofonly 6% of a wavelength.

A useful factor for characterizing antennas used for LF and Lorantransmission is the Q or quality factor. The quality factor is the ratioof energy stored in the electric and magnetic fields of the antennadivided by the energy dissipated in the radiation resistance and otherlosses over a single cycle. There is a direct relationship between Qfactor and bandwidth, as shown in equation 1 where BW is the antenna 3dB bandwidth and fc is the centre frequency.

$\begin{matrix}{{BW} = \frac{f_{c}}{Q}} & {{Eq}.\mspace{11mu} 1}\end{matrix}$

The high Q factor of typical antennas used for Loran is part of whatmakes transmission difficult and requires analysis for proper systemdesign. Impedance measurements at the antenna base can be used todetermine Q. Two antenna measurements are needed, impedance andreactance slope where reactance is the imaginary part of impedance. Theantenna can generally be modeled at a particular frequency using alumped constant series RLC circuit.

The input impedance z of the antenna equivalent circuit is shown inequation 2.

$\begin{matrix}{z = {R + {{j\omega}\; L} - \frac{j}{\omega\; C}}} & {{Eq}.\mspace{11mu} 2}\end{matrix}$

From equation 2, the reactance can be determined as the imaginary partof the impedance as shown in equation 3.

$\begin{matrix}{X = {{\omega\; L} - \frac{1}{\omega\; C}}} & {{Eq}.\mspace{11mu} 3}\end{matrix}$

The reactance slope may be determined as the derivative of the reactancewith respect to frequency as shown in equation 4.

$\begin{matrix}{\frac{\mathbb{d}X}{\mathbb{d}\omega} = {L + \frac{1}{C\;\omega^{2}}}} & {{Eq}.\mspace{11mu} 4}\end{matrix}$

Solving from Eq. 3 and Eq. 4 the inductance and capacitance of theantenna equivalent circuit can be found as shown in equations 5 and 6.

$\begin{matrix}{L = {\frac{1}{2}\left( {\frac{\mathbb{d}X}{\mathbb{d}\omega} + \frac{X}{\omega}} \right)}} & {{Eq}.\mspace{11mu} 5}\end{matrix}$

$\begin{matrix}{C = \frac{2}{{\omega^{2}\frac{\mathbb{d}X}{\mathbb{d}\omega}} - {\omega\; X}}} & {{Eq}.\mspace{11mu} 6}\end{matrix}$

Normally the antenna would be operated at resonance where the inductivereactance and capacitive reactance are equal so that the impedance ispurely resistive. For the 625′ TLM, the antenna needs to be tuned toresonance by adding an additional 39.8 μH resulting in a totalinductance of 218.8 μH in the equivalent circuit.

Once the equivalent circuit is determined, the quality factor of theantenna can be calculated using equation 7, which is true for thecircuit at its resonant frequency. From this equation it is clear thatantennas with higher resistance (radiation resistance plus losses in theconduction path including the ground return) will be lower Q and havewider bandwidth.

$\begin{matrix}{Q = {\frac{\omega\; L}{R} = \frac{1}{\omega\;{CR}}}} & {{Eq}.\mspace{11mu} 7}\end{matrix}$

Antennas used for Loran transmission are characterized by relativelyshort electrical lengths when compared to the transmitted wavelength. Atime domain representation of a Loran pulse is shown in FIG. 2.

Comparing FIG. 3A, which shows the magnitude response of a typicalnarrow-band antenna in a Loran system, and FIG. 3B which shows Loransignal pulse spectrum, it is clear that the Loran signal occupiessignificantly more bandwidth than what is available in the antenna.Objectively, the antenna 3 dB bandwidth is 1.8 kHz and the Loran pulse 3dB bandwidth is approximately 5 kHz. For most transmission systems,including AM audio broadcast in the medium wave band, the antennabandwidth generally exceeds the transmitted waveform bandwidth. Whenthere is a significant antenna bandwidth deficit, as is the case here,we can no longer think of the antenna impedance as being steady state ineither time or frequency. A transient relationship exists which requirestransient analysis techniques to be properly understood. In most allother antenna system analyses at higher frequency, steady state analysistechniques are used exclusively.

The next step is to quantify the transient effects based on the specifictransmitted waveform and the equivalent circuit that we have developed.To simplify the mathematics significantly, a frequency transformation ofthe equivalent circuit from bandpass to lowpass is used. In thistransformation, the second order antenna equivalent circuit becomes thesimplified first order equivalent circuit shown in FIG. 6.

The Loran pulse also needs to be transformed to a lowpass equivalent forthe analysis. Because the ideal bandpass pulse from equation 8 has alinear RF phase characteristic (no phase modulation), the lowpassequivalent is simply the pulse envelope shown in equation 9.i(t)=t ² e ^(−at)  Equation 9:

The lowpass equivalent circuit is governed by the network equation,equation 10.

$\begin{matrix}{{V(t)} = {{{i(t)}R} + {L\frac{\mathbb{d}i}{\mathbb{d}t}}}} & {{Equation}\mspace{14mu} 10}\end{matrix}$

Equation 11 is the result of solving for the lowpass driving voltageneeded to give the correct response.V(t)=(2Lt−aLt ² +Rt ²)e ^(−at)  Equation 11:

The lowpass equivalent circuit must have a 3 dB bandwidth which is onehalf of the 3 dB bandwidth of the antenna to achieve the same enveloperesponse. For the case of the 625′ TLM lowpass equivalent, the correctlowpass bandwidth is 909 Hz. This specific 3 dB bandwidth makes thecalculation of L and R straightforward.

FIG. 4A shows the network response.

The antenna, highly capacitive due to the short electrical length, isnormally series resonated with a loading inductor to minimize thereactance at the centre frequency. As a result of the lack of antennabandwidth, the required driving voltage envelope 400 has a verydifferent envelope than the resulting desired current envelope 402. Theinstantaneous impedance 404 of the antenna (the ratio of voltage tocurrent at one instant in time) varies throughout the pulse from a verylarge level initially to close to the steady state base impedance nearthe pulse peak and finally becoming negative during the pulse tail. Whenthe impedance is negative, power is flowing out of the antenna back tothe transmitter.

The current envelope is readily recognizable as a Loran pulse envelope.The voltage waveform is very different from the current and ischaracterized as having a high initial peak followed by a rapid decayand a negative region after the peak of the current waveform. Thedifference between the voltage and current envelope is a key concept inunderstanding the challenge of Loran transmission systems. The voltagepeak in this case is approximately five times higher than the currentpeak, representing an effective “transient voltage factor” (TVF) offive. For a system where the antenna bandwidth is greater than thesignal bandwidth (i.e., MW Broadcast), the current waveform and voltagewaveforms would be almost the same representing a TVF approaching one.In general, the TVF may be used to determine the transmitter peak powercapability, for a given antenna and pulse shape. Peak power capabilityis reduced from its ideal steady state value proportionately to the TVF.

The very high initial voltage peak shown in FIG. 4A provides the powerto rapidly increase the stored energy and current in the reactive(lossless) elements of the antenna circuit. The majority of the powerflowing from the source at the beginning of the pulse is not beingdissipated, but stored in the electric and magnetic fields of theantenna. As the current and voltage waveforms vary, the impedance seenby the transmitter varies widely through the pulse. The impedance isinitially very high, decreases to the steady state value at the currentpeak and then becomes negative in the pulse tail. A broadcasttransmitter is not designed for these variations in impedance and wouldshut down in such a situation. An important observation is that afterthe peak in the current waveform, the stored energy in the antennacircuit must be damped faster than its natural damping rate and thesource voltage waveform becomes negative as energy flows from theantenna back to the transmitter.

It is desired that the transmitters used in the present system arecapable of being modulated by digitally selecting carrier amplifiers.Typically, an exciter/modulator is placed within the system to modulatethe transmitter.

The actual architecture of the exciter/modulator is not particularlyimportant. Instead, the exciter/modulator should have the followingdesired features: 1) ability to remove failed amplifiers from themodulation sequence; 2) ability to operate with both pulsed andcontinuous modulation sequences; 3) enable and disable amplifiers in thepulsed modulation mode following a preset, prequantized sequence; 4)capability of switching the amplifiers between active and passivedamping modes, when the modulation is negative; and 5) ability toprovide an envelope quantization process for the continuous modulationprocess. The quantization block might also include a randomizer thatselects amplifiers randomly in time. The randomization serves toequalize the power delivered by individual amplifiers over time. Anotherpossible desired feature of the exciter/modulator is to include a finiteimpulse response (FIR) equalizer to equalize an external I, Q or inputfrom the continuous modulation sequence.

Transmitters for use in the present system are modulated by digitallyselecting carrier amplifiers. Typically, Class D amplifiers are used asa result of their ability to be modulated by digitally selecting carrieramplifiers. A single amplifier can only achieve very limited amplitudemodulation. As a result, a single amplifier effectively has only twostates, on or off (active or inactive). The two states only allow fortwo amplitude levels. To modulate a complex waveform, such as Loran,more amplitude states are required from the transmitter. Accordingly,some means to allow for a wide range of amplitude modulation must beemployed. This can be achieved by using many amplifiers (2 or more) andoperating them in the active or inactive states in different ratios atdifferent times (as described in Swanson, U.S. Pat. No. 4,580,111), orby using one or more amplifiers and controlling the DC power sourcevoltage (2 in FIGS. 7A-7G). It should be noted that there are numerousdifferent ways in which the source voltage could be modulated.

To achieve modulation and recover power from the antenna, each amplifiermay be operated in one of four modes: inactive mode, active mode,passive damping mode and active damping mode. The actual mode of eachamplifier can be controlled by the exciter/modulator on each full RFcycle.

For Loran, step modulation (aka digitally selecting carrier amplifiers)is used to achieve amplitude modulation as shown in FIG. 5 schematic.Step modulation makes use of the fact that a large number of amplifiersmust be combined to achieve the high power levels required. On each RFcycle or half RF cycle, the exciter makes a decision on the magnitude ofvoltage required to achieve the desired RF current waveform and thenumber of amplifiers operating is adjusted proportionately to thedesired voltage waveform. Amplifiers that are not operating on a givenRF cycle are in a low-impedance, zero voltage state. FIG. 5 shows aschematic of a step modulated transmitter using class D amplifiers.Amplifier 106 is comprised by multiple amplifiers from 1 to N (122 to126) to drive a power combiner system composed of a magnetic core andprimary winding associated with each individual amplifier, with a singlesecondary winding combining all the amplifiers outputs in series. Thecombiner output voltage is simply the sum of the input voltages,corrected for the primary to secondary turn ratio of the individualtransformers. The combined output waveform is then connected to theharmonic filter 550, antenna tuning and antenna circuit and antenna 112.It should be noted that other methods of combining may also beimplemented.

As with any transmitter system based on Class-D amplifiers, attenuationof carrier frequency harmonics is required due to the square wavevoltage produced by the amplifier at the carrier frequency. In the caseof Loran transmission, where the impedance of the antenna variessignificantly during the pulse, additional requirements are placed onthe harmonic filter design. The harmonic filter 110 is characterized byseries resonant sections in series with the antenna and parallelresonant sections in parallel with the antenna. This architecture forthe antenna provides for a low-pass filter with very low sourceimpedance and near zero phase delay such that the transmitter behaves asclosely as possible to an ideal voltage source. The behaviour of avoltage source is desirable as it has ideal transient response whichallows for maximum control over the antenna current during the widerange of impedance variations present at the input of the Loran antenna.

FIG. 6 shows a schematic of an amplifier for recovering power from anarrow-band antenna. As shown in FIG. 6, the operation of each amplifier122 to 126 is controlled by the combination of an amplifier drive logic602 and associated circuitry. The drive logic 602 and associatedcircuitry must be able to receive from the exciter at least foursignals, namely an RF drive signal 604, an enable signal 606, a dampsignal 608 and a phase signal 610, to each amplifier. The amplifierdesign enables instantaneous phase reversal as required by Loran. Thisis achieved by having an RF phase input which is the control input aswell as the separate phase input. The ability to put the amplifier into“passive damping mode” enables the antenna to free run so as to minimizelosses due to switching the amplifiers and drivers during the period oftime that power is recovered from the antenna. The drive logic enablescontrol of the transistors so that they are in any state on any cycle.

The RF drive signal 604 contains the RF phase information from the RFsource, namely the desired amplifier voltage zero crossings. For thetransmission of a Loran signal, this signal 604 would operate as acontinuous 100 kHz square wave with no phase modulation although thissignal could be any type of signal. All mode changes are madesynchronously with the RF Phase signal 610, which is used as a clock todetect the mode control signals. The amplifier itself should contain anadjustable dead time control that allows the dead time to be adjusted toallow for reduction of switching loss.

The enable signal 606 can be either inactive or active. When the enablesignal 606 is inactive, the amplifier is in a zero voltage, lowimpedance state with the top two transistors 650, 660 in the bridge inan open or off position and the bottom two transistors 670, 680 in thebridge in the closed or on position. When the enable signal 606 isactive, the amplifier may be in the Active mode, the Passive Dampingmode or the Active Damping Mode as determined by the RF drive 604, Damp608 and Phase 610 Signals.

When the damp signal 608 is active and the enable signal 606 is active,all transistors 650, 660, 670, 680 in the amplifier are shut down and itoperates as a bridge rectifier. This is the high impedance mode usedwhen the antenna impedance is negative and the power is flowing from theantenna back to the amplifier.

The phase signal 610 is used to invert the voltage output of theamplifier with respect to the RF Drive signal 604. During normal Loranoperation, the Phase signal 610 would be triggered throughout some ofthe pulses where a negative phase code pulse was desired. Additionally,if the Phase signal 610 was inverted when current was already flowing inthe antenna, the impedance phase would be inverted, i.e. positiveimpedance to negative impedance, and the amplifier mode would changefrom Active mode to Active Damping mode.

As shown in FIG. 6, the two lower transistors 670, 680 have directcoupled gate drive as their source leads are grounded. Driving thesetransistors does not require the gate drive to be floated at the B+voltage source 600. The two upper transistors 650, 660 require anisolated gate drive circuit. This isolated gate drive circuit can beachieved by using a transformer 630, 640. When the amplifier isdisabled, it is necessary that transistors 650, 660 are offcontinuously. This requires that the DC voltage be zero. To improve thetransient response of the gate drive circuitry two drivers are used foreach transformer 630, 640. This improves the gate drive transientresponse when it is started and stopped due to the amplifier beingenabled or disabled.

To achieve the inactive mode, direct DC coupling from the drive circuit602 to the transistor gates is required for the continuously ontransistors 670, 680. This DC coupling improves amplifier robustness.The improved robustness is the result of the amplifier remaining in itslowest impedance state during the presence of uncontrolled currents dueto lightning or other effects. This low impedance state is achieved whentwo transistors are on continuously and current may flow in the outputwith a low impedance. In addition, switching losses are minimized.

Since multiple amplifiers are used, a suitable series combiner isrequired. The combiner should have output current and voltagetransducers for instrumentation, protection and adaptive modulation. Inaddition, an impedance matching and harmonic filter is required. Theantenna system must be inductively loaded such that the antenna andloading inductor together form a series resonant circuit, resonant atthe transmitter carrier frequency. Accordingly, a servo controlledvariometer or some other means is required to maintain the circuit atthe correct resonant frequency. Furthermore, the loading inductorcircuit should be modified so that suitable modification of the oddharmonics will be provided.

The power required from the main DC power supply in the present systemis relatively low compared to AM and FM broadcasting. Moreover, sincethe number of amplifiers may be small, the envelope modulation system isnot readily adapted to fine regulation. As a result, a regulated B+power supply is preferred.

As a result of the high peak to average (power) ratio of the transmittedsignal, particular attention has to be paid to the type of B+ decouplingcapacitors used for the application. Moreover, controlled regulation ofthe transmitted peak pulse amplitudes is a requirement for Loran signaltransmission.

Schematic representations of the operational states of the amplifier areshown in FIGS. 7A to 7G. Gates 10, 20, 30 and 40 combined with diodes60, 70, 80 and 90 correspond to transistors 650,660, 670 and 680respectively of FIG. 6.

The active mode is characterized by the normal operation where theamplifier is contributing energy to the load or antenna 50 where thevoltage and current are in phase as shown in FIGS. 7A and 7B. Two statesmay be defined for each mode based upon the phase of the signal. Tosource energy to the antenna, the voltage applied by the amplifieracross the load 50 should be in phase with the current. When the currentis flowing in the direction of the arrow, transistors 10 and 40 areclosed and transistors 20 and 30 are open as shown in FIG. 7A. When thealternating (sinusoidal) current reverses polarity or phase, transistors10 and 40 are open and transistors 20 and 30 are closed as shown in FIG.7B. The output voltage of the transmitter is proportional to the numberof amplifiers in the active mode. The impedance recognized by theamplifier is variable depending on the current at the downstreamcombiner at that particular moment in time. The combiner current isdetermined by the number of active amplifiers as well as the load Q andthe history of modulation. The amplifier(s) remain in the active mode aslong as the load impedance remains positive and energy flow is into theantenna 50.

As shown in FIGS. 7C and 7D, in the passive damping mode, alltransistors 10, 20, 30, 40 are off and rectifiers 60, 70, 80, 90,typically in the form of diodes, associated with each transistor operateas bridge rectifiers. If MOSFET transistors are used, the rectifiers areformed by the reverse body diodes of the MOSFET and external rectifiersare not required. In certain circumstances, separate rectifiers could beused if another type of semiconductor switch is used that does not havean acceptable rectifier device. During the passive damping mode, powerflow is negative as energy (depicted by the arrow) is recovered from theantenna 50 and stored in the power supply 100 decoupling capacitors. Atthis point, the amplifier(s) essentially look like a voltage source, butwith the negative of the supply voltage and the load impedance isnegative with a level determined by the combiner current during thatparticular RF cycle and the number of amplifiers in the passive dampingmode.

In the passive damping mode shown in FIGS. 7C and 7D, the antenna iscausing current to flow in the amplifier (50 is a source of current) andall the transistors 10,20,30, 40 are open. With the transistors 10, 20,30, 40 open, the rectifiers 60, 70, 80 and 90 act as bridge rectifiersand rectified RF current flows back to the source 2 providing acapacitor 4 for storing power thus absorbing energy from the antenna 50.During one half cycle of the RF current flow, rectifiers 60 and 90 arein conduction and rectifiers 70 and 80 are off as shown in FIG. 7C.During the other half cycle, the current flow reverses, rectifiers 70and 80 are in conduction and rectifiers 60 and 90 are off as shown inFIG. 7D. When in the passive damping mode, the voltage applied to theantenna 50 is proportional to the number of amplifiers in the passivedamping mode, but in opposition to current flow making the voltageeffectively negative. At this time, the impedance seen by thetransmitter is also negative.

During passive damping the system can still be voltage modulated, sincethe output voltage of the transmitter is proportional to the number ofamplifiers that are active in the passive damping state.

In the active damping mode shown in FIGS. 7E and 7F, operation is verysimilar to the passive damping mode however, the transistors 10, 20, 30and 40 continue to switch as if they were sourcing power to the antenna50, but the phase is reversed 180 degrees such that the driving voltage,which is now inverted, is out of phase with the antenna current. Forexample in FIG. 7E gates 10 and 40 are closed, and 20 and 30 are openwhile for the reversed phase gates 10 and 40 are open and gates 20 and30 are closed. Operation is otherwise unchanged. This mode wouldnormally be activated by inverting the RF Drive signals. Typically, thismode would not be required for a standard Loran pulse, since the pulsetail phase is not critical. However, it may be beneficial to employ theactive damping mode in a case where it is necessary to achieve somedegree of phase modulation during periods where the impedance isnegative and power flow is from the antenna 50 back to the transmitter.Phase modulation may be achieved by causing the transistors 10, 20, and40 to close slightly before, or stay closed slightly longer than thenormal rectifier conduction would have occurred in the passive dampingmode. The ability to provide active damping provides control of thephase as opposed to passive damping where phase control is notmaintained. As with the passive damping mode, the system is stillvoltage modulated during active damping, since the output voltage of thetransmitter is proportional to the number of amplifiers that are activein the active damping state.

As shown in FIG. 7G, during the inactive mode of operation, theamplifier is off and is essentially out of the circuit. Both bottomtransistors 30, 40 are continuously on and both top transistors 10, 20are continuously off. Alternately, both top transistors 10, arecontinuously on and both bottom transistors 30, 40 are continuously off.Between input signals, such as Loran pulses, when no power is flowing inthe antenna 50, and when the transmitter is protecting itself fromlightning, feed line arcs or other events that threaten the RFtransistors, as determined by protection circuits and devices, allamplifiers within the transmitter are in the inactive mode. During thismode, the alternating radio frequency current from the antennacirculates from the source 50 around a loop composed of transistors 30,40 and through the ground connection between them. While this current iscirculating, the amplifier is neither contributing nor absorbing powerfrom the load as the voltage around the loop is very low.

There are limitations in the ability of the amplifier to recover powerfrom the antenna. The primary limitation is that when the antennacurrent becomes very small it is insufficient to fully charge anddischarge the self-capacitance of the transistors used in the class-Damplifier during each RF cycle. As a result the rectifiers never becomeforward biased allowing power to flow from the antenna back to the powersupply. The magnitude of RF antenna current where this effect becomessignificant is relatively small so it does not represent a significantenergy loss. However, the Loran signal specification requires that thepulse tail current be damped to very low levels at the end of the pulse.These levels are less than the level where the amplifier operates as aneffective damping system so another method of pulse damping is requiredlate in the pulse tail. In the preferred embodiment, one or moreamplifiers are modified to operate only as a resistive damping elementwhen required at the end of the pulse. The amplifier is modified byremoving the connection to the power supply 2 and the top transistors 60and 70 while adding an appropriate resistor in parallel with the load50. The resistive damping element operates in one of two modes. In thefirst mode, the inactive mode, transistors 80 and 90 are on and loadcurrent circulates from the load through transistors 80 and 90 and theamplifier is effectively out of the circuit. In the active modetransistors 80 and 90 are open and the antenna current flows through theresistor in parallel with the load. In the active mode the resistivedamping element effectively places the damping resistor in series withthe load current thus damping the antenna current without thelimitations described when an amplifier recovers power from the antenna.

In the low-pass equivalent analysis, as seen in FIG. 4A, the voltagesource needs to be able to operate with a negative voltage after aboutthe 80 microsecond point in the pulse. Similarly, the voltage also needsto be negative for a transmitter operating on the radio frequency signalat 100 kHz. This is achieved when the voltage waveform is inverted andthe RF current and voltage at the transmitter output are out of phase.The voltage phase reversal can be seen in FIG. 8.

The advantage of damping the antenna current in the way described aboveis that the amplifiers effectively operate as rectifiers allowingantenna current to flow back to the DC power supply, recovering theenergy stored in the antenna. This significantly increases the overallsystem efficiency. A transmitter operated without the capability torecover power suffers from an additional efficiency factor related tothe energy in the pulse tail that is wasted. FIG. 4B shows the powerflow in the antenna system derived from the voltage and currentwaveforms where the power flow is simply the product of the voltage andcurrent envelopes. The antenna with the higher Q, more narrow bandwidth,is less efficient in delivering power and therefore will benefit morefrom the ability to recover stored power.

The efficiency limit calculation is shown in equation 12 where ηMAX isthe upper bound of efficiency, ES is the energy flowing from thetransmitter to the antenna calculated from the area under the positivepower portion of the curve and ER is the energy flowing from the antennato the transmitter calculated from the area under the negative powerflow portion of the curve.

$\begin{matrix}{\eta_{MAX} = \frac{E_{S} - E_{R}}{E_{S}}} & {{Eq}.\mspace{11mu} 12}\end{matrix}$

Without power recovery capability, the absolute efficiency limit for anantenna Q of 55 is 67% and for an antenna Q of 120 the limit is 37%.Another way of looking at this is the increase in AC power consumptionof the transmitter. For a Q of 55 the transmitter without power recoverycapability will consume 49% more power (all other efficiency factorsequal) and for an antenna Q of 120 the transmitter without powerrecovery capability will consume 2.7 times more power. It should benoted that this efficiency factor is solely based on the ability torecover stored energy in the antenna and is not related to otherefficiency factors such as the DC to RF conversion efficiency of the RFamplifier or the AC to DC conversion efficiency of the main powersupply.

The Class D amplifiers using MOSFET switching transistors have veryshort turn on and turn off times on the order of a few tens ofnanoseconds. The switching times are due to fixed inductances andcapacitances associated with the semiconductor and its package. Theswitching times are very repeatable allowing the transmitter to have avery constant delay time from pulse trigger input to resulting RF outputwaveform. Pulse to pulse jitter times should be on the order of 1nanosecond or less.

Because of the high efficiency of the system, power supply design issomewhat simplified. The use of a tightly regulated primary power supplyas well as ample energy storage capacitors allow for very tight pulse topulse amplitude regulation. For the prototype system, the theoreticalpulse amplitude regulation through a nine pulse group is 0.3%.

Loran antennas themselves may be somewhat inefficient with the antennaefficiency of typical North American systems in the range of 50 to 75%.This factor must also be included in overall system efficiency.

The average radiated power of a typical Loran system is surprisinglylow. Radiated power may be calculated based on the energy of an idealpulse. Integration of Equation 8 yields a pulse energy of 83.4μJoules/watt peak which is the total energy in a single pulse of 1 rmswatt on the peak half cycle. So a system which radiates 400 kW peak isradiating 33.4 Joules per pulse. For a single rated 400 kW systemrunning at 125 pulses per second, the total radiated power is only 4.2kW. However for an antenna that is only 50% efficient, the transmittermust deliver 8.4 kW. The total power consumed, based on a 70%transmitter efficiency would be 12 kW. This power would scale up to 29kW at 300 pps.

FIG. 9 shows a method of operating an amplifier for providing powerrecovery from a narrow-band antenna. A signal source is provided at 910.The signal state is determined at 912 either based upon characteristicsof the signal source or by a predetermined switching sequence, as in thegeneration of a Loran pulse. The impedance state of the antenna isdetermined at 914. Either the antenna current is being amplified ordampened. If amplification is occurring, the impedance state of theantenna is positive and active mode from one or more amplifiers isrequired. The phase of the signal is determined at 916. Based upon thephase of the signal a first state A of transistor configuration, such asshown in FIG. 7A, is selected and enabled at 918. When the phase changesto state B a second state as shown in FIG. 7B is selected at 920. Thefirst and second states can be alternated based upon a positive ornegative phase of the signal. In addition the amplifiers alternatebetween states sequentially. For example for a positive phase theamplifier would switch between state 7A and 7B, where as for a negativephase it would switch between 7B and 7A. If amplification is notoccurring, the impedance state of the antenna is negative and dampingmode from one or more amplifiers is required at 922. Damping mode canoccur in either an active mode or passive mode based upon signal dampingrequirements. If passive mode is required either a first of secondpassive damping mode is selected at 930 as shown in FIG. 7C or 7D. Ifactive damping is required at 922, the phase of the signal is determinedat 924. Based upon the phase of the signal a first active damping stateA of transistor configuration, such as shown in FIG. 7E, is selected andenabled at 926. When the phase changes to state B a second state asshown in FIG. 7F is selected at 928. During damping current flows to thevoltage source charging one or more capacitors and thus recoveringpower. As in the active mode, the states of the damping modes may bealternated between the two respective state configurations based uponthe initial phase state for passive and active damping. At each stage aspecific number of amplifiers may be utilized proportional to thedesired magnitude of the voltage waveform. If the desired impedance ispositive the selected amplifiers are in the driving mode, if the desiredimpedance is negative the selected amplifiers are in one of the twodamping modes. It is assumed that amplifiers that are not in use are ina disabled state, such as FIG. 7G and not contributing to amplification(positive impedance) or damping or recovering power (negativeimpedance).

In amplification module 106 multiple amplifiers are utilized to generatethe required signal. For example to generate a Loran pulse, a 15amplifier configuration may be utilized. The amplifiers are switched tooperating states sequentially to generate the pulse. To generate aspecific wave form such as the Loran pulse the amplifiers would beswitched in a predefined sequence such as:

8, 14, 15, 13, 9, 6, 3, 1, −1, −2, −2, −3, −3, −3, −3, −2, −2, −2, −2,−1, −1, −1, −1, −1 and −1

In descriptive form the sequence is represented as:

-   -   Pulse Trigger    -   Enable 8 amplifiers driving positive phase then negative phase    -   Enable 14 amplifiers driving positive phase then negative phase    -   Enable 15 amplifiers driving positive phase then negative phase    -   Enable 13 amplifiers driving positive phase then negative phase    -   Enable 9 amplifiers driving positive phase then negative phase    -   Enable 6 amplifiers driving positive phase then negative phase    -   Enable 3 amplifiers driving positive phase then negative phase    -   Enable 1 amplifier driving positive phase then negative phase    -   Enable 1 amplifier passive damping for a full cycle    -   Enable 2 amplifiers passive damping for a full cycle    -   Enable 3 amplifiers passive damping for a full cycle    -   Repeat 3 times    -   Enable 2 amplifiers passive damping for a full cycle    -   Repeat 3 times    -   Enable 1 amplifier passive damping for a full cycle    -   Repeat 5 times.

Note that this voltage envelope has been “sampled” once per full cycleinterval i.e. the envelope sampling frequency is equal to the carrierfrequency. However sampling at half cycle may be utilized to have finercontrol of the pulse shape. This would mean that the envelope samplingfrequency is two times the carrier frequency.

This would change the sequence for the first 2 RF cycles of the abovefor example to:

-   -   Pulse Trigger    -   Enable 8 Amplifiers driving positive phase    -   Enable 11 Amplifiers driving negative phase    -   Enable 14 Amplifiers driving positive phase    -   Enable 15 Amplifiers driving negative phase

By switching the state of the amplifier and the number of amplifierscomplex waveforms can be generated. The switching may occur in apredefined sequence or based upon the characteristics of an inputsignal.

A typical transmitter source voltage waveform (at the combiner output)is shown in FIG. 8. This particular sequence is matched to an antenna Qof 55. The number of amplifiers enabled on sequential RF cycles are 8,14, 15, 13, 9, 6, 3, 1, −1, −2, −2, −3, −3, −3, −3, −2, −2, −2, −2, −1,−1, −1, −1, −1 and −1. In this sequence, positive numbers indicate thenumber of amplifiers sourcing power and negative numbers indicate thenumber of amplifiers recovering power. The remaining amplifiers would bedisabled such as shown in FIG. 7G. The transmitter voltage may bemodulated in the damping mode by switching amplifiers proportionatelyfrom the damping mode to disabled mode. The transmitter still hascontrol of the output voltage waveform even when recovering power fromthe antenna as the output voltage will be proportional to the number ofamplifiers in the damping mode on a given RF cycle.

It should be noted that the present disclosure can be carried out as amethod, can be embodied in a system, a computer readable memory forcontrolling one or more amplifiers.

It will be understood that numerous modifications thereto will appear tothose skilled in the art. Accordingly, the above description andaccompanying drawings should be taken as illustrative of the inventionand not in a limiting sense. It will further be understood that it isintended to cover any variations, uses, or adaptations of the inventionfollowing, in general, the principles of the invention and includingsuch departures from the present disclosure as come within known orcustomary practice within the art to which the invention pertains and asmay be applied to the essential features herein before set forth, and asfollows in the scope of the appended claims.

1. An amplitude modulated amplifier for amplifying a radio frequency(RF) signal, the amplifier comprising: four selectably controllabletransistors arranged to receive power from a direct current (DC) powersupply to amplify the signal, the transistors configured in a bridgeconfiguration with a first and a second transistor coupled between theDC power supply and an antenna having a short electrical length comparedto a wavelength of a transmitted signal, and with a third and a fourthtransistor coupled between the antenna and ground in an ‘H’configuration, each transistor having a gate input to determine thestate of the transistor between an on state and an off state; fourdiodes each associated with a different transistor of the fourtransistors and forward biased from a source to a drain of eachtransistor, forming a bridge rectifier; and drive logic for controllingeach of the four transistors from an open state to a closed amplifyingstate based on external input, wherein the drive logic is selectablefrom a driving state to a damping state of the amplifier by switchingtransistors to recover power from the antenna when an impedance state ofthe antenna is negative, and to deliver the recovered power to the DCpower supply.
 2. The amplifier of claim 1 wherein the transistors are ina first driving state when the first and fourth transistors are on andthe second and third transistors are off to provide current to theantenna in a positive phase of a voltage waveform through thetransistors, and the transistors are in a second driving state when thesecond and third transistors are on and the first and fourth transistorsare off to provide current to the antenna in a negative phase of avoltage waveform through the transistors.
 3. The amplifier of claim 2wherein the transistors are in a passive damping mode when the first,second, third and fourth transistors are off to enable current to flowin reverse from the antenna to the DC power source through the diodesassociated with the transistors for the positive and negative phases ofthe voltage waveform.
 4. The amplifier of claim 3 wherein thetransistors are in a first active damping state when the first andfourth transistors are switched on and the second and third transistorsare switched off when impedance at the antenna is negative to enablecurrent to flow in reverse from the antenna to the DC power sourcethrough the transistor and diode associated with each transistor toprovide current to the DC power source in the positive phase of thevoltage waveform; and the transistors are in a second active dampingstate when the second and third transistors are switched on and thefirst and fourth transistors are switched off to enable current to flowin reverse from the antenna to the DC power source through thetransistor and diode associated with each transistor to provide currentto the DC power source in the negative phase of the voltage waveform;wherein each active damping state maintains phase control over thevoltage waveform during damping.
 5. The amplifier of claim 1 whereineach transistor has a low off state current when a large voltage isacross a switch thereof to reduce off state losses and allow highervoltages to be used to increase power capability.
 6. The amplifier ofclaim 1 wherein each transistor has a low device capacitance andinductance, and is adapted to change state quickly.
 7. The amplifier ofclaim 1 wherein each transistor comprises an integral reverse diodehaving low voltage and resistance during reverse current flow as well aslow current with off state voltage.
 8. The amplifier of claim 1 whereineach transistor comprises a Metal Oxide Semiconductor Field EffectTransistor (MOSFET) device including a reverse body diode.
 9. A systemfor transmitting radio frequency signals into an antenna having a shortelectrical length compared to a wavelength of a transmitted signal, thesystem comprising: a signal source providing a baseband source signal tobe amplified and modulated before transmission; an exciter module forup-converting the source signal based on a frequency of a received RFsource; an amplification module comprising one or more amplifiers, eachamplifier comprising: four selectably controllable transistors arrangedto receive power from a direct current (DC) power supply to amplify theup-converted source signal, the transistors configured in a bridgeconfiguration with a first and a second transistor coupled between theDC power supply and an antenna, and a third and fourth transistorscoupled between the antenna and ground in an ‘H’ configuration, eachtransistor having a gate input to determine the state of the transistorbetween an on state and an off state; four diodes each associated with adifferent transistor of the four transistors and forward biased from asource to a drain of each transistor, forming a bridge rectifier; drivelogic for controlling each of the four transistors from an open state toa closed amplifying state based on external input, wherein the drivelogic is selectable from an active state to a damping state of theamplifier by switching transistors to recover power from the antennawhen an impedance state of the antenna is negative, and to deliver therecovered power to the DC power supply; a power source for delivering DCpower to the amplification module or receiving and storing power fromthe amplification module; an antenna coupled to the amplificationmodule, the antenna having a highly capacitive characteristic whereinthe signal wavelength is less than the antenna wavelength, whereincurrent is delivered from the amplification module to the antenna whenin the driving state and impedance is positive, and current is receivedfrom the antenna and delivered to the DC power source in the dampingstate when impedance is negative; and a controller coupled to orintegrated with the exciter module, for controlling the operation of theone or more amplifiers via the drive logic, wherein a subset of the oneor more amplifiers is switched between the driving state or dampingstate, wherein any remaining amplifier(s) of the one or more amplifiersare placed in an inactive mode, to control the modulation of thetransmitted signal.
 10. The system of claim 9 where the one or moreamplifiers are switched between driving state, damping state, orinactive state in a predefined sequence on sequential RF cycles or halfcycles to modulate the signal from the signal source to generate a Loranpulse.